Paired ZF Sampling for Pulse Running Time Filling Level Sensor

ABSTRACT

A pulse running time filling level sensor includes a sampling device for sampling an IF signal at discrete instants and for converting the sampling values into digital sampling values, and a digital signal processing device for subsequent processing of the digital sampling values by calculating at least one new value characterizing the IF curve from respectively exactly two digital sampling values.

REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. patent application Ser. No.11/625,946 filed Jan. 23, 2007 which claims the benefit of the filingdate of German Patent Application Serial No. 10 2006 006 572.7 filedFeb. 13, 2006 and U.S. Provisional Patent Application Ser. No.60/772,701 filed Feb. 13, 2006. The disclosure of the abovepatents/applications is hereby incorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to level measuring. In particular, thepresent invention relates to a method for level measuring for a pulserunning time filling level sensor, and a level measuring instrument,which determines the filling level of a filling material in a containeraccording to the pulse running time method.

BACKGROUND INFORMATION

For continuous filling level determination in containers, containinge.g. liquids or bulk materials, frequently sensors are used, whichmeasure according to the pulse running time method the running time ofelectromagnetic or sound waves from sensor to filling material surfaceand back. From the interval, determined from the pulse running time viathe wave propagation speed, between sensor and filling material surface,if the sensor's location of installation with respect to the containerbottom is known, the desired filling level may be calculated directly.

Sound waves may be generated and radiated by so-called ultrasoundfilling level sensors approximately in the range from 10 kHz to 100 kHzby means of electromechanical sound transducers. The reflected soundpulses are either received by the same sound transducer, or by a secondtransducer, provided only for receiving, and evaluated regarding theirrunning time with respect to the time of transmission.

Electromagnetic waves, which may be located in a frequency range betweenapproximately 0.5 and 100 GHz, are radiated and received again by thesensor, usually via antennas. In addition, instruments are known, whichreturn the wave along a wave guide from the sensor to the fillingmaterial. The reflection of the waves at the filling material surface isdue to the variation of the propagation impedance for the wave at thispoint.

The pulse radar method takes advantage of the generation of shortcoherent microwave pulses, so-called bursts, and determines the directperiod between transmission and receipt of the pulses. For ordinarymeasuring distances, ranging up to several meters, the time intervals tobe measured are extremely short, which is the reason why with pulseradar sensors, the echo signal received may be appropriately retardedthrough a time transformation method. Such a method is described in DE31 07 444. It supplies a retarded echo signal corresponding to the highfrequency transmit and receive signal received, but which is sloweddown, e.g. by a factor of between 10,000 and 100,000. A carrier wavefrequency of the microwave pulse of e.g. 5.8 GHz is transformed into acarrier wave frequency of the retarded echo pulse between e.g. 58 kHzand 580 kHz. This signal, created internally by time transformation, isin general also called intermediate frequency signal, or abbreviated asIF signal, and is usually situated between 10 kHz and 1 MHz, e.g.between 50 kHz and 200 kHz. As mentioned before, this IF signal is aretarded image of the time course of the transmitted and receivedmicrowave pulses. Both regarding the frequency range and the nature ofthe amplitude course, the IF signal of the pulse radar method and theecho signal of the ultrasound method are similar, which is the reasonwhy further processing and evaluation of these signals for determiningthe relevant echo running time, and thus the measuring distance, may bethe same except for minor differences. Therefore, if in the subsequentdescription, IF signals are mentioned, these are meant to imply not onlythe retarded representations of the microwave signals, but also theultrasound echo signals having basically the same aspect.

An IF signal contains a time course of individual pulses, starting witha reference pulse or echo derived from the transmit pulse, via variouspulses or echoes from reflection points in the propagation path of thewaves, where the impedance of the propagation medium changes. Each pulseconsists of a carrier wave of a certain fixed frequency with a pulseshaped amplitude course defined by the shape of the transmit pulse. Thetotality of all echoes for a given time between the occurrence of thereference echo and the maximum required running time for a relevantmeasuring range makes up the IF signal. A measuring cycle of a fillinglevel sensor in question is characterized by the formation of at leastpart of an IF signal, but usually one or more complete IF signals, andsubsequent signal processing, evaluation, measurement value formation,and measurement value output, making use of the IF signal formed.Periodical repetition of measuring cycles may ensure updating ofmeasurement values in order to follow up on variable filling levels.

In order to separate in a possibly occurring variety of echoes withinone IF signal the one echo from the filling material surface fromadditionally occurring clutter, the individual echoes have to berecognized by means of characteristic features. One important feature isthe course of the amplitude of an echo, with an amplitude rise at thebeginning, a maximum amplitude, and an amplitude decay at the end of theecho. This amplitude course is obtained by forming the envelope of theIF signal. In envelope formation, the information on the carrier wavephase course of the echoes is usually lost. However, as taking advantageof knowing the phase course may allow for a significant increase inmeasuring accuracy, methods are known, wherein in addition to mereenvelope information, also phase information of an IF signal isevaluated.

The desired running time of the echo from the filling material surfaceresults from the time interval between reference echo and fillingmaterial echo. This may be determined from the interval of twocharacteristic points of the envelope, e.g. the peak interval of bothechoes or of envelope points on the echo edge, which have a definedamplitude relation with the peak. With phase information, this runningtime information derived from the envelope may be corrected, resultingin higher accuracy.

An example of such two-part signal processing and evaluation of the IFsignal may be found in DE 44 07 369. The level measuring instrumentdescribed therein comprises an analog signal processing channel forforming the envelope, and a channel parallel thereto with an analogquadrature demodulator for the IF signal for generating a firstquadrature signal representing the real part of the IF signal and asecond quadrature signal representing the imaginary part. Due to theanalog construction of both channels, certain problems caused bycomponent tolerances and long term drift appear in signal processing,which may lead to a decrease in measuring accuracy.

Furthermore, it has to be noted that the amplitude differences betweenechoes of well reflecting surfaces at dose range and poorly reflectingsurfaces at the end of the measurement range may be very large.Amplitude differences of more than 120 dB, corresponding to a stressratio of 1 to one million, may appear, and may have to be handled by thesensor's signal processing. If for envelope formation, the common methodof half or full wave rectification is used, e.g. via analog diodecircuits, with subsequent low-pass filtering, such a dynamic range canhardly be managed. In order to alleviate the requirements, an IF signalamplifier may be implemented with variable amplification adapted to theecho running time. This amplification control or STC (sensitivity timecontrol) may reduce the required dynamic range of all subsequent stagesof signal processing. Alternatively, it may be possible to vary theamplification gradually within the IF signal, or between differentsuccessive IF signals. Thereafter, the amplitude information of theindividual stages with reduced dynamic range may be summed up toinformation with full dynamic range. The logarithmic processing of thesignal for compressing the envelope amplitudes may be another method. Anexample of such signal processing with a hardware logarithmizer, whichat the time of logarithmizing also performs signal rectification, andthus, together with subsequent low-pass filtering, may allow for thelogarithmic envelope to be formed, is found in DE 101 64 030. Thisdocument also shows a solution as to how the phase information can alsobe obtained from the logarithmized signal, and thus the dual processingof amplitude and phase may be minimized.

In order to avoid the disadvantages of largely analog signal processing,e.g. long term drift, component tolerances, and lack of flexibility withrespect to variable sensor parameters, a mostly digital processing ofthe IF signal is to be aimed at. For this purpose, it may be advisableto sample the IF signal, upon possible analog signal amplification andlow-pass or band-pass filtering, in order to avoid aliasing, and toconvert the time discrete sampling values into a digital valuerepresenting the voltage value. This method is called A/D conversion. Adigitally stored sampling sequence represents the analog IF signal withechoes contained therein. Both amplitude and phase information of the IFsignal is preserved, and available for subsequent digital signalprocessing. However, the requirements regarding the necessary samplingfrequency, the amplitude resolution of A/D conversion, and the memoryand computing load of digital signal processing may be problematic.Therefore, analog signal processing and logarithmic envelope formationmay be combined with IF digitizing. The echo amplitudes are evaluatedfrom the logarithmic envelope, eventually also digitized, whereas fromthe digitized IF signal, only additional phase information may have tobe derived. Thereby, IF digitizing may be simplified in that it can berestricted exclusively to the two time ranges in the signal containingthe reference echo and the relevant echo from the filling materialsurface. This may save memory space, computing time, and once IFamplification has been adjusted, also amplitude resolution of the A/Dconverter. However, on the analog side in turn, more circuit means maybe required.

Another method of digitally sampling the IF curve is known from DE 10140 346. It features a relatively low sampling frequency and easyenvelope formation, but may require considerable synchronization meansfor sampling the IF waves exactly at the peaks of the carrier frequency.

SUMMARY OF THE INVENTION

According to an exemplary embodiment of the present invention, a methodis proposed, wherein the IF signal is sampled at discrete points in time(i.e. discrete instants), and converted into a digital value, whereinthe subsequent processing of the digital values calculates fromrespectively exactly two digital sampling values adjacent in time atleast one new value characterizing the IF curve.

This may lead to a low sampling frequency, low memory means, lowcomputational cost for subsequent digital processing regarding theevaluation of amplitude and phase information, flexible adaptation tovariable sensor parameters, high signal sensitivity, and high measuringaccuracy.

According to another sample embodiment of the present invention, thisvalue is an envelope value, or a value describing the phase.

It appears that from only two sampling points of the IF signal,approximately one envelope value as well as one phase value may becalculated, if the time interval between the sampling points is known,and certain time intervals are avoided. With an adequate choice of thesampling interval, the calculated envelope value may be so close to theactual value that the remaining error is negligibly small.

According to a sample embodiment of the invention, the remaining errormay be further decreased by an iterative approximation method, if theslope of the envelope at the point to be calculated is known. The slopein turn may be estimated very well from the course of several adjacentenvelope points calculated by approximation.

According to a sample embodiment of the inventive method, two samplingvalues consecutive in time always form a sampling pair, wherein the timeinterval ta1 between the two points of a pair is less than half theinterval ta2 between consecutive sampling pairs.

Furthermore, the time interval between the two points of a pair may bechosen so as to be smaller than half a period of the IF carrier wave.All time intervals of the two points of a sampling pair corresponding tohalf a period of the IF, or a multiple thereof, are to be avoided at anyrate. Also, the sampling interval for a pair of points may be controlledin a precise manner so that it corresponds to a quarter of the period ofthe IF carrier frequency. However, otherwise the interval may be definedas about one eighth of the IF period.

According to a sample embodiment of the invention, the time interval ta2between consecutive sampling pairs may be chosen to be so large that allof the sampling and digitizing of the IF signal now no longer satisfiesthe Nyquist sampling theorem. The basis therefore is the limitedbandwidth of the IF signal resulting from the pulse shaped nature of theecho signals contained therein. Such under-sampling allows for certainsavings in terms of power consumption of A/D conversion, memoryrequirements, and computing time.

In accordance with a first configuration, depending on the IF frequencyf_(IF) and IF bandwidth B, the time intervals ta2 of the consecutivesampling pairs are chosen according to the following aspects: Theduration must be less than the reciprocal value of twice the IFbandwidth, i.e. ta2<1/(2*β), and the sampling duration must not bewithin the range between ta2=n/(2*f_(IF)+B), and ta2=n/(2*f_(IF)−B),wherein n=1, 2, 3, . . . .

There are so-called stop bands, wherein sampling aliasing effects mayalter the signal. However, all other intervals between the samplingpairs may be chosen, whereby a method thus implemented in a sensor maybe adapted very flexibly to various requirements.

In order to avoid undesirable DC components in the sampled IF signal, itmay be appropriate, to place the interval ta2 between the sampling pairsapproximately half-way between two forbidden areas. Then, throughdigital band-pass filtering of all digital values of an IF curve, bothan interfering DC component and other interfering signals, e.g. noise,close to the IF frequency band can be eliminated. Such digital band-passfiltering can be applied e.g. separately to respectively the group ofall first sampling values of all pairs and the group of all secondvalues of all pairs of the IF curve.

Alternatively to the choice, which as just been described, of the timeinterval between the sampling pairs, this may also be exactly the periodof the IF carrier frequency or an even-numbered multiple thereof. Inthis case, noise components, but no DC components, of the IF signal maybe filtered out digitally. In this case, instead of a digital band-passfilter, a digital low-pass filter may be sufficient. In addition, acoherent system averaging of the sampled IF values may be performed. Theterm system averaging designates a formation of the mean value overseveral values corresponding in time of different consecutive IF curves.

According to a sample embodiment of the present invention, the twodigital sampling values are adjacent in time.

According to a sample embodiment of the present invention, the timeinterval of both digital sampling values is one quarter of the IFfrequency period.

According to a sample embodiment of the present invention, the envelopeis formed as the root of a sum of squares of the two digital samplingvalues.

According to a sample embodiment of the present invention, the timeinterval of the two digital sampling values is one eighth of the IFfrequency period.

According to another sample embodiment of the present invention, thedigital sampling values are assigned alternately to one of two groups,wherein the characteristic value is calculated from respectively onevalue of each group.

Prior to computing, digital filtering of each group, or coherent systemaveraging of each group can be performed.

The time intervals of the digital sampling values of each group can besmaller than the reciprocal value of a double bandwidth.

According to another sample embodiment of the present invention, thetime intervals of the digital sampling values of each group are greaterthan the reciprocal value of a double frequency, so that they don'tsatisfy the sampling theorem.

According to another sample embodiment of the present invention, for thetime intervals of the digital sampling values of each group, thefollowing holds true:

${{{Time}\mspace{14mu} {intervals}} \notin \left\lbrack {\frac{n}{{2*f_{IF}} + B},\frac{n}{{2*f_{IF}} - B}} \right\rbrack},$

-   -   with f_(IF): IF carrier frequency        -   bandwidth of the IF signal        -   n: natural number, n=1, 2, 3, . . . .

According to another sample embodiment of the present invention, thetime intervals of the digital sampling values of each group are situatedapproximately half-way between two stop bands.

${{Time}\mspace{14mu} {intervals}} \approx \frac{{2*f_{IF}*\left( {{2*n} + 1} \right)} - B}{{8*f_{IF}^{2}} - {2*B^{2}}}$

Furthermore, digital band filtering of each group with a centerfrequency of about one quarter of the sampling frequency may beperformed.

The time intervals of the digital sampling values of each group maycorrespond to the IF frequency period or a multiple thereof.

Also, digital low-pass filtering of each group may be performed.

The envelope value HK_(i) and the phase value φ_(i) may be calculatedfrom the two sampling values of one group respectively according to

${HK}_{i} = \sqrt{{{IF}\; 1_{1}^{2}} + \frac{\left( {{{IF}\; 2_{i}} - {{IF}\; 1_{i}*{\cos \left( {\omega*\left( {{ta}\; 1_{i}} \right)} \right)}}} \right)^{2}}{{\sin \left( {\omega*\left( {{ta}\; 1_{i}} \right)} \right)}^{2}}}$${\tan \left( \phi_{i} \right)} = \frac{{{\sin \left( {\omega*t\; 2_{i}} \right)}*{IF}\; 1_{i}} - {{\sin \left( {\omega*t\; 1_{i}} \right)}*{IF}\; 2_{i}}}{{{\sin \left( {\omega*t\; 1_{i}} \right)}*{IF}\; 2_{i}} - {{\cos \left( {\omega*t\; 2_{i}} \right)}*{IF}\; 1_{i}}}$

wherein:

-   -   i: running variable, i=0, 1, 2, . . .    -   HK_(i): envelope value    -   φ_(i): phase value    -   ta1 _(i)=t2 _(i)−t1 _(i); time interval between sampling of IF1        _(i) and IF2 _(i)    -   IF1 _(i): sampling value from the first group    -   t1 _(i): sampling instant of IF1 _(i)    -   IF2 _(i): sampling value from the second group    -   t2 _(i): sampling instant of IF2 _(i)    -   ω: =2*π*f_(IF); angular frequency of the IF carrier wave

According to another sample embodiment of the present invention, thephase value is computed from sampling values and envelope values.

According to another sample embodiment of the present invention, thetime interval between two consecutive sampling pairs is greater thanhalf the period of the IF signal.

According to another sample embodiment of the present invention, theinterval between two consecutive sampling pairs does not match exactlyone period.

According to another sample embodiment of the present invention, for thetime interval ta2 between two consecutive sampling pairs, the followingfurther holds true:

${{ta}\; 2} \notin \left\lbrack {\frac{n}{{2*f_{IF}} + B},\frac{n}{{2*f_{IF}} - B}} \right\rbrack$

-   -   with f_(IF): IF carrier frequency        -   B: bandwidth of the IF signal        -   n: natural number, n=1, 2, 3, . . . .

According to another sample embodiment of the present invention, a pulserunning time filling level sensor for determining a filling level in acontainer is proposed, the filling level sensor comprising a samplingdevice for sampling an IF signal at discrete points in time (or timeinstants), and for converting sampling values into digital samplingvalues, and a digital signal processing device for subsequent processingof the digital sampling values through calculation of at least one newvalue characterizing the IF curve from respectively exactly two digitalsampling values.

Hereafter, with reference to the figures, preferred sample embodimentsof the present invention will be described.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a functional diagram of a radar filling level sensor.

FIG. 2 shows a signal processing flow-chart of a signal processing witha quadrature demodulator for evaluating envelope and phase.

FIG. 3 shows a functional diagram of a radar filling level sensor forcarrying out the inventive method.

FIG. 4a and FIG. 4b show a signal processing flow-chart of the signalprocessing according to various embodiments of the invention.

FIG. 5 shows an ideal IF signal.

FIG. 6 shows a detail of the IF signal of FIG. 5 with marking of IFsamplings according to the state of the art.

FIG. 7 shows a detail of the IF signal of FIG. 5 with marking of IFsamplings according to an exemplary embodiment of the inventive method.

FIG. 8 shows errors in the envelope calculation according to anexemplary embodiment of the inventive method before and afterapplication of the improved calculation method.

FIG. 9 shows a signal processing graph of analog and digital signalprocessing according to an exemplary embodiment of the invention.

FIG. 10 shows a sample illustration of an IF signal, sampling points,envelope, and phase angle when applying an exemplary embodiment of theinventive method.

The illustrations in the figures are schematic and not to scale.

In the following description of the figures, the same reference symbolswill be used for identical or similar items.

DETAILED DESCRIPTION

FIG. 1 shows a radar filling level sensor for determining a fillinglevel in, for example, a tank 1001, operating according to the pulserunning time method, with a high frequency circuit section 10, a powerunit 20, analog signal processing 30, an A/D converter 40, and amicrocontroller 50 with peripherals, e.g. memory 52, display 54, clockand reset circuit 53. The sensor is supplied for instance via thetwo-wire circuit 21, which also outputs the filling level value inanalog form, applied by the power source 22 and the controller 56, as acurrent of 4 . . . 20 mA. On the supply line 21, digital communicationof the sensor with the outside may also take place simultaneously.Materially, this may be made possible by means of the digital interface55. As is known to those skilled in the art, the power unit contains aDC/DC converter 23 with series-connected memory capacitor 24 forsupplying all other circuit sections.

The high frequency circuit 10 contains a transmit generator 11 forgenerating high frequency transmit pulses with a predefined pulserepetition frequency. They are directed via the directional coupler 12,or alternatively via a circulator, to the antenna 13, and radiatedtherefrom. A given part of the transmit pulses directly reaches thereceiving amplifier 14 via the directional coupler 12, and forms a timereference stamp, or more simply put a reference echo. At a later time,the echo from the reflection of the transmit pulse sent at the fillingmaterial surface, and possibly echoes of other existing obstacles in thepropagation path of the microwaves, will follow. These echoes arereceived by the antenna 13, and directed via the directional coupler 12to the receiving amplifier 14. Both the reference echo and the otherechoes reach the sampler or mixer 15, after amplification. Therein, theyare sampled with a second pulse train, which is generated in thesampling generator 16. As a result of this sampling, an IF signal 17 iscreated, which contains in retarded form all echoes of the highfrequency signal of the receiving amplifier 14. In this respect,reference is made to DE 31 074 44, which describes the operation of thisretarding method in detail. A very schematic, idealized IF signal 17 isshown in FIG. 5. For this example, a radar frequency of 24 GHz and aretarding factor of 150,000 were chosen. Thereby, an IF carrier wave of160 kHz is created. The propagation constant of the microwaves,resulting from light speed, of 3.33 ns/m is retarded to 0.5 ms/m. Due tothe round trip, a time difference of Ims may thus be created between twoechoes, the mutual interval thereof being Im. The echoes of an IF signalare plotted at this time scale. In the idealized IF signal of FIG. 5,for simplicity's sake, only two echoes with a mutual interval of 0.5 ms,respectively of 0.5 m, are plotted. The echo 171 is the reference echo,whereas the echo 172 is the reflection from the filling materialsurface. As already mentioned, real IF signals often contain furtherclutter, making signal evaluation more complicated. The duration of theIF signal depends on the sensor's measuring range of interest. In FIG.5, for improved illustration of the IF wave, the measuring range waschosen very short.

Each echo of the IF signal consists of an amplitude modulated carrierwave with the IF frequency. The desired interval information may bedetermined from the amplitude course, and possibly also from the phasecourse.

In order to obtain this information, according to the method representedin FIG. 1, or the circuit represented in FIG. 1, the IF signal 17 may beprocessed by analog signal processing 30, and then, converted intodiscrete digital values with the analog-digital converter 40. Thesedigital values are saved in memory 52, so that the microcontroller 50can have access thereto.

In the represented embodiment, the processing of the analog signal ismainly performed separately by amplitude and phase information. Foramplitude information, so far the implementation of a hardwarelogarithmizer 33 has come to mind, which may rectify and logarithmizethe IF signal. For adapting the IF signal 17 to the level range thereof,the amplifier 31 is used, whereas the band-pass filter 32 improves thedetectability even of the smallest echoes by filtering out, in as far aspossible, all clutter outside the frequency band of the IF frequency.

The low-pass filter 34 after the logarithmizer smoothes the envelopecreated by eliminating IF frequency components. If the amplitude dynamicrange in the IF signal exceeds the dynamic range of the logarithmizer33, then the IF amplifier 31 may be varied regarding the amplificationthereof. This variation of amplification can take place gradually orcontinuously for the duration of an IF signal, or else stepwise from oneIF signal to the next. If the variation of amplification applied isknown, the digitized logarithmic envelops thus created can be correctedand grouped by the microcontroller 50 so that correct amplitudeinformation for all echoes is available.

The signal path with the IF amplifier 35 and the band-pass filter 36serves to obtain phase information. The IF amplifier 35 adapts the IFsignal in amplitude to the voltage range of the A/D converter 40, andthe band-pass filter 36 also serves both to filter out noise componentsand as an anti-aliasing filter for the subsequent analog-digitalconversion.

When transitioning from analog to digital level, the Nyquist samplingtheorem may have to be noted, which indicates that the samplingfrequency must be at least twice as high as the highest frequencyoccurring in the signal. In order to satisfy this condition, on the onehand, the analog low-pass or band-pass filtering of the IF signal beforesampling, and on the other hand, rather large-scale oversampling may beadvisable, which in the above mentioned example leads to samplingfrequencies of over 500 kHz. Knowing, as already mentioned, that theamplitude dynamic range in the IF signal can be very large, an A/Dconverter with high amplitude resolution, i.e. with high bit width, hasto be chosen. However, such relatively fast A/D converters, which at thesame time offer high resolution, are hardly available in the light ofthe additional aspects of lower power consumption and reasonable cost.In view of reducing the required amplitude resolution of the converter,of course, as described above, in the formation of the analog envelope,an analog IF amplifier 35 with STC functionality, or an amplifier withstepwise amplification may be provided, too. In case of stepwiseamplification, several digitized IF signals of variable amplificationmay be combined mathematically into a complete IF signal with highdynamic range. The same applies for the stepwise switching of theamplification within an IF signal. Of course, it may also be possible touse instead of an IF amplifier variable in amplification, severalamplifiers of parallel construction of variable amplification, thesignals of which are converted by the A/D converter almost at the sametime, or else consecutively. However, at any rate, it is to be notedthat for the benefit of a reduced amplitude resolution of the A/Dconverter, higher demands are made in terms of sampling frequency,memory requirement and/or load or processing speed of themicrocontroller 50 for digital signal processing.

A detail of an IF signal sampled according to the Nyquist samplingtheorem is shown in FIG. 6. The first half of the echo 172 of FIG. 5 isshown with the original IF curve 17 and the equidistant sampling pointsin the time interval ta, from which a digital value is formed.Furthermore, a dashed line shows the envelope. The sampling interval tarepresented in this example is 1.95 us corresponding to a samplingfrequency of 512 kHz, which is about three times the IF carrierfrequency of 160 kHz.

For the digital signal processing of these sampling values in the signalprocessing software 51, the implementation of the method of quadraturedemodulation is appropriate, which is represented schematically in FIG.2. The IF signal values, sampled and digitized by the A/D converter 40,which are available in memory 52, can be filtered, if required, in theband-pass filter 511 in order to improve the signal/noise ratio. Forthis purpose, digital filter algorithms known to those skilled in theart, like FIR (finite impulse response) and IIR (infinite impulseresponse) filters, are suitable. Further improvement of the signal/noiseratio can be obtained, if required, by system averaging or predetectionintegration 512. This is a mean value formation of the sampling valuesof various consecutive IF signals. Those values are averagedrespectively, which were sampled at the same relative time intervalswith respect to the beginning of the IF signal.

The IF sampling values thus filtered are then multiplied in themultiplier 514 with a mathematically generated sine signal 513 of thesame frequency. This corresponds to frequency conversion to the carrierfrequency of zero. Higher frequency components created at the same timeare filtered out by the digital low-pass filter 515. In parallel, the IFsignal values are multiplied in the multiplier 519 with a mathematicalcosine signal 518, resulting from the 90° shift of the sine in the phaseshifter 517, and also filtered in the low-pass filter 520. The inphase(I) 516 and quadrature (Q) 521 components thus calculated are the basisfor calculating the phase (522), and possibly also the envelope (523).Here, the phase is calculated from the arc tangent of the quotient,phase=arctan(Q/I), the envelope from the root of the sum of squares,envelope=SQR(Î2+Q̂2).

In comparison with an analog version, this digital quadraturedemodulation may be independent from component tolerances, and easy toadapt to variable conditions. However, through digitizing according tothe sampling theorem, relatively many digital values are created, towhich relatively many arithmetic operations according to the diagram ofFIG. 2 may have to be applied. This may result in relatively high memoryrequirements and high computing load for the microcontroller 50, whichleads to compromises in as far as manufacturing cost, electricalconsumption, and cycle time of measurement value updating is concerned,which are far from ideal.

Globally, it appears from FIGS. 1 and 2 and the explanations thereofthat the analog and digital signal processing of IF signals describedtherein is very demanding in as far as computing time, electricalconsumption, and cost is involved, if the measurement values output bythe filling level sensor are to be very accurate.

In contrast, FIGS. 3 and 4 show sample embodiments for analog anddigital signal processing according to the present invention.

Herein, FIG. 3 differs from FIG. 1 only regarding the analog signalprocessing 30′ and different digital signal processing 57. The branchfor analog formation of the logarithmic envelope was omitted in thisexample, because according to the invention, the envelope may becalculated very easily from the sampled digital values of the IF signal.

For this purpose, as represented in FIG. 4a , the sampling values of anIF signal are divided into two groups IF1 (521) and IF2 (522). In thesimplest case, this is done by assigning all IF values consecutivelysampled by the A/D converter at the sampling time intervals ta1 701 andta2 702 alternately to one of both groups. It has appeared herein thatfrom a value pair 571, consisting respectively of one sampling valuefrom each group, a corresponding envelope value 572 and/or phase value573 can be calculated for the IF signal. It is assumed that the IFfrequency f_(IF) is well known. Furthermore, the sampling instants ofall sampling values must of course be known, wherein not absolute, butrelative instants with respect to a beginning instant of the IF signalare relevant. With a given sampling raster, the instant may e.g. resultsimply from the memory address where the value was stored.

The formulae of calculation for the individual values i of the envelopeand the phase are:

$\begin{matrix}{{HK}_{i} = \sqrt{{{IF}\; 1_{1}^{2}} + \frac{\left( {{{IF}\; 2_{i}} - {{IF}\; 1_{i}*{\cos \left( {\omega*\left( {{ta}\; 1_{i}} \right)} \right)}}} \right)^{2}}{{\sin \left( {\omega*\left( {{ta}\; 1_{i}} \right)} \right)}^{2}}}} & (1) \\{{\tan \left( \phi_{i} \right)} = \frac{{{\sin \left( {\omega*t\; 2_{i}} \right)}*{IF}\; 1_{i}} - {{\sin \left( {\omega*t\; 1_{i}} \right)}*{IF}\; 2_{i}}}{{{\sin \left( {\omega*t\; 1_{i}} \right)}*{IF}\; 2_{i}} - {{\cos \left( {\omega*t\; 2_{i}} \right)}*{IF}\; 1_{i}}}} & (2)\end{matrix}$

wherein: i: running variable, i=0, 1, 2, . . .

-   -   HK_(i): envelope value    -   φ_(i): phase value    -   ta1 _(i)=t2 _(i)−t1 _(i); time interval between sampling of IF1        _(i) and IF2 _(i)    -   IF1 _(i): sampling value from the first group    -   t1 _(i): sampling instant of IF1 _(i)

IF2 i: sampling value from the second group

-   -   t2 _(i): sampling instant of IF2 _(i)    -   ω: =2*π*f_(IF); angular frequency of the IF carrier wave

These calculated values may not yield with mathematical accuracy thedesired amplitudes of the envelope and phase values. An error may occur,which depends on the variation of the envelope between the two points ofa pair. The bigger the envelope variation, the greater the possibleerror. Therefore, the time interval between the two sampling instants ofone pair may be kept relatively small. FIG. 7 shows for the same timedetail of the IF signal as in FIG. 6 a favorable sampling raster. Twosamplings in the time interval of ta1 701 are performed cyclically,wherein the first value is assigned to the first group, and the secondone to the second group. This is repeated for time interval ta2 702. Therespective sampling values are plotted as points on the IF signal. Theenvelope values calculated from each sampling pair according to theformula above can be seen in FIG. 7 as solid squares on the idealenvelope plotted with a dashed line. In this example, for the timeinterval ta1, a time corresponding to one quarter of the IF frequencyperiod was chosen. During this time, the envelope value only changesnegligibly, so that the calculation error created is relatively small.The choice of one quarter of the period is favorable because in thiscase, π/2 results from the calculation of the envelope according to theformula (1) as the arguments of the sine and cosine functions appearingtherein. Thereby, calculating the formation of the root from the sum ofsquares of the two sampling values is simplified:

HK _(i)=√{square root over (IF1_(i) ² +IF2_(i) ²)}  (3)

Furthermore, it is to be noted that in addition to intervals ta1 to bechosen advantageously, it may also be necessary to avoid certainintervals. These are all time intervals ta1 corresponding to half of theIF frequency period, and all multiples thereof.

Another very favorable time interval ta1 701 has proved to be the onecorresponding approximately to one eighth of the IF frequency period, asduring this very short time, the envelope variation, and thus thecalculation error, is very small.

Another or additional way for reducing the error results from amathematically enhanced calculation approach, wherein the amount ofenvelope variation ΔHK_(i) between the two sampling points of a pair hasto be determined in addition. Herein, an iterative approximation methodis appropriate, wherein first the still error prone envelope values arecalculated according to the formula (1) above. From the envelope valuesHK_(i−1) and HK_(i+1) adjacent to a given envelope point, theapproximate slope of the envelope is calculated from the straight linethrough both adjacent points, and therefrom the first approximation ofthe envelope variation ΔHK_(i). Then, the following iterativecalculation is applied for improved determination HK1 of this givenenvelope value HK_(i):

$\begin{matrix}{{{{HK}\; 1_{j + 1}} = {{{HK}\; 1_{j}} - \frac{{a\; 4*{HK}\; 1_{j}^{4}} + {a\; 3*{HK}\; 1_{j}^{3}} + {a\; 2*{HK}\; 1_{j}^{2}} + {a\; 1*{HK}\; 1_{j}} + {a\; 0}}{{4*a\; 4*{HK}\; 1_{j}^{3}} + {3*a\; 3*{HK}\; 1_{j}^{2}} + {2*a\; 2*{HK}\; 1_{j}} + {a\; 1}}}}\mspace{79mu} {with}\mspace{79mu} {{j = 0},1,2,\ldots}\mspace{79mu} {{a\; 4} = {\sin^{2}\left( {\omega*{ta}\; 1_{i}} \right)}}\mspace{79mu} {{a\; 3} = {2*{\sin^{2}\left( {\omega*{ta}\; 1_{j}} \right)}*\Delta \; {HK}_{i}}}{{a\; 2} = {{{\sin^{2}\left( {\omega*{ta}\; 1_{i}} \right)}*\Delta \; {HK}_{i}^{2}} + {2*{\cos \left( {\omega*{ta}\; 1_{i}} \right)}*{ZF}\; 1_{i}*{ZF}\; 2_{i}} - {{ZF}\; 1_{i}^{2}} - {{ZF}\; 2_{i}^{2}}}}\mspace{79mu} {{a\; 1} = {2*{ZF}\; 1_{i}*\Delta \; {HK}_{i}*\left( {{{\cos \left( {\omega*{ta}\; 1_{i}} \right)}*{ZF}\; 2_{i}} - {{ZF}\; 1_{i}}} \right)}}\mspace{79mu} {{a\; 0} = {{- {ZF}}\; 1_{i}^{2}*\Delta \; {HK}_{i}^{2}}}\mspace{79mu} {and}\mspace{79mu} {{{HK}\; 1_{0}} = {HK}_{i}}\mspace{79mu} {{\Delta \; {HK}_{i}} = \frac{\left( {{HK}_{i + 1} - {HK}_{i - 1}} \right)*{ta}\; 1_{i}}{{t\; 1_{i + 1}} - {t\; 1_{i - 1}}}}} & (4)\end{matrix}$

This iterative approximation method for solving a 4th-orderNewton-Raphson equation may be abandoned after any number of steps, andmay result in an improvement of the calculated envelope values.

FIG. 8 shows the absolute calculation error (calculated value of theenvelope minus exact value of the envelope) for the envelope of the IFsignal of FIG. 5 when sampling according to the example of FIG. 7 as adashed line before, and as a solid line after application of 5iterations of the approximation method, plotted for the time scale. Theassociated maximum amplitudes are 1 for the reference echo and 0.4 forthe filling material echo. It may be seen therefrom that the error isreduced from a maximum of about 2.5% of the correct envelope value tovalues of a maximum of about 0.1%.

The phase value, which by definition is the phase angle of the carrierwave of a given echo at an instant once defined, can alternatively tothe formula of calculation (2) above, wherein just as for the envelopecalculation, some error is created, be determined from the much moreaccurate envelope value according to the application of theapproximation method. For this purpose, the IF wave of an echo is to beconsidered as a complex index, of which the amount (=envelope) andimaginary component (=sampling value) are known at a current instant.Thereby, a current phase angle at this instant can be calculated. Thedesired phase angle of the echo at the previously defined instant simplyresults from the time interval between the current and defined instantsand the period of the IF carrier wave.

The basic signal processing sequence of this preferred embodiment isrepresented in FIG. 4b . It is different from the sequence in FIG. 4aonly in that in another arithmetic operation 574 the previouslydetermined envelope value is improved by the approximation methoddescribed. If required, in another calculation step 575, an exact phasevalue is determined from the improved envelope value. Of course, asplotted with a dashed line, the phase value may also be derived from theenvelope value calculated initially, if application of the approximationmethod is omitted.

If sampling according to the inventive example of FIG. 7 is comparedwith sampling of FIG. 6, it is noticed that when applying the inventivemethod it is now no longer mandatory to satisfy the Nyquist samplingtheorem. The interval ta2 702 between the sampling pairs can be chosenfreely within certain limits. These limits only result from the IFcarrier frequency, and the bandwidth of the IF signal. As it is known,the sampling of a signal can be considered as a multiplication of thesignal with a Dirac pulse train. When considered in the frequency range,this corresponds to a convolution of the frequency spectrum of the IFsignal with the Fourier transform of the Dirac pulse train. The resultof this convolution are a great (in theory infinite) number of frequencybands with IF bandwidth. The repetition frequency of the Dirac pulsetrain, which determines the sampling interval ta2 702, is responsiblefor the situation of these frequency bands. For the sampled signal toremain unaltered, the frequency bands must not partially overlap. Thisresults in the following requirements for the time interval ta2 betweenthe sampling pairs:

$\begin{matrix}{{{ta}\; 2} \leq \frac{1}{2*B}} & (5) \\{{{{ta}\; 2} \neq \frac{n}{{2*f_{IF}} + B}},\frac{n}{{2*f_{IF}} - B}} & (6)\end{matrix}$

with f_(IF): IF carrier frequency

-   -   B: bandwidth of the IF signal    -   n: natural number, n=1, 2, 3, . . . .

For the example used so far, with f_(IF)=160 kHz and a supposedbandwidth B=20 kHz, it follows that the time interval ta2 should besmaller than 25 ρs. Furthermore, the time interval ta2 must for instancenot be situated in the range between 2.94 ρs and 3.33 μs. The abovementioned inequality results in 7 other stop bands for ta2 for n=2 to 8,which are not all mentioned explicitly herein. When choosing theappropriate sampling interval ta2, preferably a value about half-waybetween two stop bands is chosen according to the formula

${{ta}\; 2} \approx \frac{{2*f_{IF}*\left( {{2*n} + 1} \right)} - B}{{8*f_{IF}^{2}} - {2*B^{2}}}$

With reference to the numerical values mentioned, this is e.g. aninterval of about 10.9 μs as represented in FIG. 7, or 7.7 μs or 4.6 μs.

The IF frequency component, which is relevant for further signalprocessing, is situated for the preferred selection of the samplinginterval at about one quarter of the sampling frequency. For the exampleabove with ta2=10.9 μs the midband of the IF band involved is at about22.9 kHz, the bandwidth being once again 20 kHz. This means thatfiltering can be applied to the sampling sequences, separated into bothgroups. In this case, this could be done with a band-pass filter havinga center frequency of 22.9 kHz and a bandwidth of 20 kHz. Byimplementing such a digital filter, the embodiment of which as a FIR orIIR filter is known by those skilled in the art, both noise components,which have not been suppressed by the analog band-pass filter before A/Dconversion, and an interfering DC component in the IF signal can befiltered out efficiently.

In contrast with the just described definition of the sampling intervalta2 between the adjacent sampling pairs, it is also possible to chooseexactly an interval according to one period of the IF carrier frequency,or of an integer multiple thereof. This has the advantage that the IFsignal is reproduced accurately into the baseband, and that instead ofthe band-pass filter, a digital low-pass filter can be used. However,herein a possible DC component on the IF signal can no longer befiltered out, and the formation of the envelope can thus be altered.

In addition to digital band-pass or low-pass filtering, system averagingof the sampled values also comes to mind. As already mentioned, systemaveraging is the formation of a mean value for sampling values ofdifferent consecutive IF signals, corresponding to the same echo runningtime. If system averaging is performed before formation of the envelope,then it is coherent system averaging, or else predetection integration,for system averaging of the envelope values it is incoherent systemaveraging or postdetection integration. Although both methods suppressnoise with respect to the echo signal, coherent system averaging isclearly more efficient in this respect. Coherent system averaging can beapplied very easily according to the inventive method by separate systemaveraging of the two sampling groups. In comparison with incoherentsystem averaging, which can of course be carried out alternatively oradditionally with the calculated envelope values, the computationalmeans are merely doubled.

In FIG. 9, a configuration of analog and digital signal processing for apulse running time filling level sensor is represented schematically.This version takes into account high signal dynamics in that the IFsignal is split into two parallel channels, in which the IF amplifiers35 a and 35 b with different amplifications v1 and v2 are provided.Accordingly, there are also two analog band-pass filters 36 a and 36 b,and two A/D converters 40 a and 40 b. It is supposed that the resolutionof an A/D converter is not sufficient for digitizing both the largestand the smallest echoes. However, if this were possible, then of coursea one-channel construction would be sufficient. According to the diagramrepresented in FIG. 7, each A/D converter converts the sampling valuesof the IF signal into respectively two groups IF1 . . . and IF2 . . . ,which are saved in memory 52 for further digital processing withindigital signal processing 57. It is to be noted here in addition thatthe function of the two A/D converters can also be supported by a singleA/D converter provided with at least two inputs. The two groups ofdigital values, respectively, of which the groups IF11 521 and IF21 522stand for instance for small echoes and the groups IF12 523 and IF22 524for large echoes, in a relation of v1>v2, are digitally filteredgroupwise if required 576 a, 576 b, 577 a, 577 b, and coherently systemaveraged 578 a, 578 b, 579 a, 579 b, as described above. This comes tomind at any rate for the groups of small echoes, as therein echoes maybe are partially covered by noise. Thereafter, according to theinvention, pairs are formed 571 a, 571 b, from which the envelope valuescan be calculated 572 a, 572 b according to the formula (1) or (3)above. After incoherent system averaging 580 a, 580 b, if required, thetwo separate envelops for large and small echoes are combined into onecomplete envelope 581. For this purpose, all that is required is theknowledge of the difference of amplification between v1 and v2 as wellas of some overlap of the two dynamic ranges.

This complete envelope 582 is the basis for the identification 583 ofthe reference echo and the filling material echo from the totality ofechoes of a curve train. The small amplitude errors described above inenvelope calculation can be tolerated for this evaluation step withoutany problem.

For the identification 583 of the two desired echoes, e.g. the followingprocessing steps are required: First of all, from the whole IF signal,individual echoes have to be separated and characterized as to theirfeatures. Echo detection resorts e.g. to a threshold value curve, whichis either permanently predefined for the instrument, or generateddynamically during operation. The envelope values saved are comparedregarding their amplitude with the time dependent threshold. Envelopevalues situated above the threshold indicate an echo at this location.So as to detect only the relatively largest echoes of an IF signal, itis possible to lower the threshold progressively, until a sufficientamount of echoes has been detected. Detected echoes are characterizedfor features like e.g. maximum amplitude, noise ratio, echo length, echoshape, mean value, center of gravity, and edge steepness. From theseecho features, it is also possible to some extent to draw conclusions asto the type of container, the nature of the filling material surface,and the environmental conditions. Furthermore, from the mutual echointervals, certain reflection conditions inside the container, such asobliquely positioned bulk material surfaces, or multiple reflectingcurved container covers, can be derived. This information can help onthe one hand to select application-specific software parameters in viewof improved processing of the sampled echo values, and on the otherhand, to provide the user, in addition to mere filling levelinformation, also with additional indications on environmentalconditions in the container, functional security of the sensor, andmeasuring in general.

From the totality of the echoes of an IF signal, by means of allfeatures collected, the reference echo and the echo from the fillingmaterial surface are finally identified. In particular, theidentification of the filling material echo may not be so easy. Methods,which in addition to the mere features of the echoes resulting from thegiven IF signal also evaluate the history of a measurement, are knownfrom U.S. patent application Ser. No. 11/202,007 (now U.S. Pat. No.7,333,900) or U.S. Prov. Patent Application Ser. No. 60/601,929, towhich reference is made herein.

In order to reach a measurement value for the filling level, the timeinterval between the reference echo and the filling material echoidentified has now to be determined 584. First of all, this measurement584 resorts to the digital envelope values of both echoes. However,knowing that they, as mentioned already several times, contain smallcalculation errors, for a limited amount of envelope points, theapproximation method according to the formula (4) above can be applied.The relevant points are for instance those of the envelope in the timerange of the two echoes to be measured. The information on their timeposition is transmitted from the block for identification 583 to theblock for improving the envelope values after the approximation method574. For the same points, also a calculation of the phase 575 can beperformed so as to further increase the measuring accuracy. As to how tointegrate phase values for improvement of the measuring accuracy,reference is made herein to DE 44 07 369.

FIG. 10 shows individual curve courses or sampling and calculationvalues from the signal processing chain of FIG. 9. At the top, theidealized analog IF signal, and in addition sampling values takentherefrom as points may be seen. The representation is limited to one ofthe two channels. At the bottom, the envelope values calculated fromrespectively one sampling pair, and the phase values derived therefromin the unit rad may be seen.

When measuring the echo interval 584 on the basis of the envelopevalues, it appears that the interval between the respective peaksdescribes the echo interval only very inaccurately. Clearly betterresults are provided by the measurement between points at the echoedges, the amplitude of which have a defined relation with therespective maximum amplitude of the echo. Further improvement isobtained by interpolation of intermediate points of the sampled envelopepoints in the area of the echo edges to be measured. Suitableinterpolation methods, such as linear interpolation, polynomialinterpolation, or spline interpolation, are generally known.Alternatively, there are also methods, wherein the position of an echois determined in that a previously defined standard echo or a portionthereof is made coincident, as far as mathematically possible, with thepoints of the echoes currently to be measured. From the position thusdefined of the standard echo, the desired time position of the currentecho results. Methods for correlating the IF curve or the envelope witha so-called standard echo pursue the same objective. At the instant atwhich the correlation result is at a maximum, the standard echo bestfits the curve involved, which is the reason why at the same time thisinstant is also representative for the time position of the echo.Eventually, for each measuring cycle, a measurement value 585 for thedistance of the filling material echo from the reference echo iscreated. If the defined sensor and container parameters are known, thisvalue may be converted into a filling level.

Finally, further improvement of the measuring accuracy may be obtainedin that such measurement values 585 of various consecutive measuringcycles are averaged 586. The time constant of the averaging may be setpermanently, or else, it may be adapted dynamically. If newly determinedmeasurement values are only slightly different from the measurementvalue determined previously, the time constant is increased to someextent. However, if several new measurement values one after the otherare sufficiently different therefrom, this indicates filling materialmovement, and the time constant should be reduced. Likewise, in someinstances, it can be useful to provide a hysteresis for the measurementvalue output. This is justified if it has to be assumed that thedirection of movement of the filling material surface is inverted onlyrarely, and in this case, a slight delay of the measurement valuesoutput is tolerated. The measurement value thus defined 587 is finallythe one analogically and/or digitally output by the sensor as thefilling level.

It has thus been shown that by sampling relatively few values of the IFsignal, the envelope and also the phase angle of the IF signal may becalculated very easily, from respectively only two sampling points.Thereby, the need for memory space and computing time for digitizing andevaluating the IF signal may be considerably reduced, without giving upthe advantages offered by mainly digital signal processing. Theseadvantages reside in that one is independent from component tolerances,that processing is easily adapted to variable sensor parameters, that itmay be possible to obtain a signal/noise ratio via digital filtering andcoherent or incoherent system averaging, and that measuring accuracy maybe increased through phase evaluation. Furthermore, the samplingfrequency may be adapted to different sensor types and different powersupplies of the sensors. For two-wire sensors with analog measurementvalue output as current picked up between 4 and 20 mA, the poweravailable to the sensor is relatively small. This condition can be copedwith by adapting the sampling frequency. It may even be possible to varythe sampling frequency within an IF curve, e.g. in order to sample therelevant areas of reference echo and filling material echo more finelythan the remaining areas without the echoes to be measured.

The method may be suitable both for replacing and supplementing theanalog logarithmizer mainly used so far with consecutive digitizing ofthe logarithmic envelope.

The implementation of the invention is not limited to the preferredembodiments represented in the figures. Rather, a plurality of variantscan be envisaged, which make use of the represented solution and theinventive concept, even in case of fundamentally different types ofembodiments.

Additionally, it is to be noted that “comprising” does not exclude anyother items or steps, and that “a” or “an” do not exclude a plurality.Furthermore, it is to be noted that features or steps having beendescribed with reference to one of the above sample embodiments can alsobe used in combination with other features or steps of other embodimentsdescribed above. Reference numerals in the claims are not to beconstrued as limitations.

1-27. (canceled)
 28. A method for digitizing and subsequent processingof an intermediate frequency signal for level measuring using a pulserunning time filling level sensor, comprising: pair sampling theintermediate frequency (IF) signal at pairs of discrete points in time,wherein the intermediate frequency signal comprises an amplitudemodulated carrier wave having an intermediate frequency; converting thesampled IF signal into sampling pairs of digital sampling values; andcalculating from respectively exactly one of the sampling pairs at leastone characterizing value characterizing the IF signal; wherein for atime interval between two consecutive sampling pairs the following holdstrue: a length of the time interval is less than 1/(2*B) and is notwithin a finite number of one or more pre-defined stopbands, whereineach of the one or more pre-defined stopbands is bounded by a lower anda higher limit, respectively, and each of the one or more stopbandscovers a respective time-range between, respectively, the lower and theupper limit, wherein the respective lower limit is defined by$\frac{n}{{2*f_{IF}} + B},$  and wherein the respective higher limit isdefined by $\frac{n}{{2*f_{IF}} - B}$ wherein each respective lowerlimit of the respective stop band is less than 1/(2*B), with: f_(IF): IFcarrier frequency; B: bandwidth of the IF signal; and n: all positiveintegers up to the largest positive integer so that the respective lowerlimit of the respective stop band is less than 1/(2*B).
 29. The methodaccording to claim 28, wherein the two digital sampling values withineach pair are adjacent in time.
 30. The method according to claim 29,wherein the time interval of the two digital sampling values is onequarter of an IF frequency period.
 31. The method according to claim 29,wherein the time interval of the two digital sampling values is oneeighth of an IF frequency period.
 32. The method according to claim 28,wherein the digital sampling values are assigned alternately to one oftwo groups, wherein each sampling pair comprises the first digitalsampling value and the second digital sampling value, wherein one of thetwo groups is defined by the respective first sampling value from eachsampling pair and wherein the other one of the two groups is defined bythe respective second sampling value from each sampling pair.
 33. Themethod according to claim 32, wherein the characterizing value iscalculated from respectively one value of each group.
 34. The methodaccording to claim 32, further comprising: before the calculating step,digital filtering each group.
 35. The method according to claim 32,further comprising: before the calculating step, coherent systemaveraging each group.
 36. The method according to claim 34, whereindigital band filtering of each group with a center frequency of aboutone quarter of the sampling frequency is performed.
 37. The methodaccording to claim 32, wherein the time intervals of the digitalsampling values of each group correspond to the IF frequency period or amultiple thereof.
 38. The method according to claim 37, furthercomprising: digital low-pass filtering each group.
 39. The methodaccording to claim 28, wherein the characterizing value is one of anenvelope value and a phase value.
 40. The method according to claim 39,wherein the at least one characterizing value is an envelope value,wherein the calculation of the envelope value for the respectivesampling pairs is performed according to${HK}_{i} = \sqrt{{{IF}\; 1_{1}^{2}} + \frac{\left( {{{IF}\; 2_{i}} - {{IF}\; 1_{i}*{\cos \left( {\omega*\left( {{ta}\; 1_{i}} \right)} \right)}}} \right)^{2}}{{\sin \left( {\omega*\left( {{ta}\; 1_{i}} \right)} \right)}^{2}}}$wherein: i: a non-negative integer, with i indicating respective ones ofthe sampling pairs; HK_(i) is an i-th envelope value for a respective,i-th, one of the sampling pairs; IF1 _(i): a respective first digitalsampling value of a respective, i-th, one of the sampling pairs; IF2_(i): a respective second digital sampling value of a respective, i-th,one of the sampling pairs; ta1 _(i): time interval between sampling ofIF1 ₁ and IF2 i; and ω: angular frequency of the carrier wave.
 41. Themethod according to claim 39, further comprising: iteratively improvingthe envelope value based on an approximation method.
 42. The methodaccording to claim 39, further comprising: calculating a phase valuefrom the sampling values and envelope values.
 43. The method accordingto claim 28, wherein the time interval between two consecutive samplingpairs does not match exactly one period of the IF signal.
 44. The methodaccording to claim 32, wherein time intervals of the digital samplingvalues of each group do not satisfy the Nyquist sampling theorem so thatthe time intervals are larger than a reciprocal value of twice the IFfrequency.
 45. A pulse running time filling level sensor for digitizingand subsequent processing of an intermediate frequency signal fordetermining a filling level in a tank, comprising: a sampling devicepair sampling the intermediate frequency (IF) signal at pairs ofdiscrete points in time, the sampling device converting sampling valuesinto sampling pairs of digital sampling values, wherein the intermediatefrequency signal comprises an amplitude modulated carrier wave having anintermediate frequency; and a digital signal processing devicesubsequently processing of the digital sampling values throughcalculation of at least one characterizing value characterizing the IFsignal from respectively exactly one of the digital sampling pairs, andwherein for a time interval between two consecutive sampling pairs thefollowing holds true: a length of the time interval is less than 1/(2*B)and is not within a finite number of one or more pre-defined stop bands,wherein each of the one or more pre-defined stop bands is bounded by alower and a higher limit, respectively, and each of the one or more stopbands covers a respective time-range between, respectively, the lowerand the upper limit, wherein the respective lower limit is defined by$\frac{n}{{2*f_{IF}} + B},$  and wherein  the respective higher limit isdefined by $\frac{n}{{2*f_{IF}} - B},$ wherein each respective lowerlimit of the respective stop band n is less than 1/(2*B), with: f_(IF):IF carrier frequency; B: bandwidth of the IF signal; and n: all positiveintegers up to the largest positive integer so that the respective lowerlimit of the respective stop band is less than 1/(2*B).
 46. The sensoraccording to claim 45, wherein the two digital sampling values withineach pair are adjacent in time.
 47. The sensor according to claim 46,wherein the time interval of the two digital sampling values is onequarter of an IF frequency period.
 48. The sensor according to claim 45,wherein the characterizing value is one of an envelope value and a phasevalue.
 49. A pulse running time filling level sensor for digitizing andsubsequent processing of an intermediate frequency signal fordetermining a filling level in a tank, comprising: a sampling devicepair sampling the intermediate frequency (IF) signal at pairs ofdiscrete points in time, the sampling device converting sampling valuesinto sampling pairs of digital sampling values, wherein the intermediatefrequency signal comprises an amplitude modulated carrier wave having anintermediate frequency; and a digital signal processing devicesubsequently processing of the digital sampling values throughcalculation of at least one characterizing value characterizing the IFsignal from respectively exactly one of the digital sampling pairs, andwherein for a time interval between two consecutive sampling pairs thefollowing holds true: a length of the time interval is less than 1/(2*B)and is not within a finite number of one or more pre-defined stopbands,wherein each of the one or more pre-defined stopbands is bounded by alower and a higher limit, respectively, and each of the one or morestopbands covers a respective time-range between, respectively, thelower and the upper limit, wherein the respective lower limit is definedby $\frac{n}{{2*f_{IF}} + B},$  and wherein the respective higher limitis defined by $\frac{n}{{2*f_{IF}} - B}$ wherein eachn<N=(2*f_(IF)+B)/2*B, with: n: a positive integer; f_(IF): IF carrierfrequency; B: bandwidth of the IF signal; and n: all positive integersless than N.